Measurement system including resonant bridge, and phase and amplitude sensitive discriminator



G. W. COOK Nov. 16, 1965 MEASUREMENT SYSTEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISCRIMINATOR Filed OCb. 7, 1959 9 Sheets-Sheet l IN VEN TOR. G50/@65 M cva/f M, my

G. W. COOK Nov. 16, 1965 MEASUREMENT SYSTEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISCRIMINATOR Filed 001;. 7, 1959 9 Sheets-Sheet 2 WJS ...WN

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MEASUREMENT SYSTEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISGRIMINATOR Filed Oct. 7, 1959 9 Sheets-Sheet 5 A/H- GAP MASS 0N E/vo CAPM/70H5 0F CANT/LEVER A/n GAP cAPAc/rons\ 45 40 /22/1 IN VEN TOR. E0/P65 M coo/f 147- 7 OPA/E K5 G. W. COOK Nov. 16, 1965 MEASUR`E1MENT SYSTEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISCRIMINATOR Filed OCT.. '7, 1959 9 Sheets-Sheet 4 Nov. 16, 1965 G, w, COOK 3,218,551

MEASUREMENT SYSTEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISCRIMINATOR Filed Oct. 7, 1959 9 Sheets-Sheet 5 Meg 2 M 2 M M L :L L. l-

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MEASUREMENT SYSTEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISCRIMINATOR Filed Oct. '7, 1959 l 9 Sheets-Sheet 6 F 7,4 RECEIVER G. W. COOK MEASUREMENT SYSTEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISCRIMINATOR 9 Sheets-Sheet '7 Filed 00T.. '7. 1959 Nov. 16, 1965 G, w, COOK 3,218,551

MEASUREMENT SYSTEM INCLUDING REsoNANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISCRIMINATOR Filed Oct. 7, 1959 9 Sheets-Sheet 8 INVEN TOR. Ea/Pf M a0/1 www Nov. 16, 1965 G. w. cooK 3,218,551

MEASUREMENT SYSTEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSITIVE DISCRIMINATOR Filed Oct. 7, 1959 9 Sheets-Sheet 9 Amcos wc f cos wmf Am cos (wmf +7r) Am 00S wm f RELATIVE AMPL/TUDE CA RR/E R PHA SE /A/ VE RS/ON FREQUENCY //V K/LOCYCLES AMPL/TUDE /N PERCENT 0F MAX/MUM m, MMM

United States Patent O 3,218,551 MEASUREMENT SYSIEM INCLUDING RESONANT BRIDGE, AND PHASE AND AMPLITUDE SENSI- 'IiVE DISCRIMINATR George W. Cook, Washington, DI., assigner to Thioirol Chemical Corporation, Bristol, Pa., a corporation of Delaware Filed Get. 7, 1959, Ser. No. 844,993 Claims. (Cl. 324-61) The present invention relates to a measurement system including a resonant bridge and associated equipment for providing accurate measurement of various conditions or phenomena. In particular, the measurement system of the present invention provides accurate quantitative measurement of small changes in electrical capacitance for measuring various conditions or phenomena such as force, displacement, pressure, velocity, torque, temperature and the like. The system is quick and sensitive in its response to changes in capacitance and conveniently provides an output voltage which varies linearly with changes in the condition under observation.

The resonant bridge circuit includes two arms formed by a pair of capacitors which diiferentially vary in accordance with the condition or phenomenon under observation. The other two arms of the bridge are formed by electrical impedances which are not physically present within the bridge circuit but which are reected into the bridge circuit from other portions of the system outside of the bridge circuit. In other words, two arms of the bridge circuit are phantom arms which are not physically present but which are electrically present in the circuit and are in operation, providing advantageous results.

In the illustrative embodiment of the present invention described herein, there is provided an improved phase and amplitude-sensitive discriminator coupled to the output of the bridge circuit. This discriminator includes four half-wave voltage doublers which advantageously provides a reference to the common return or ground every half cycle and thus avoid any tendency for the output voltage to drift. Moreover, this discriminator is resonant with both the signal input circuit and the reference input circuit, and thus the only energization required for the discriminator is that small amount which is necessary to sustain oscillation therein.

In prior practice, measuring systems have been devised utilizing electrical capacitors as sensitive elements, but prior systems have suffered from the disadvantages of one or more of the following arrangements:

1) The distributed capacitance of the cable extending from the sensitive capacitor ele-ment to the indicating circuit was connected in parallel with the sensing capacitance. In this arrangement the sensitivity was reduced and changes in cable capacitance introduced spurious signals and indications which were relatively large and which masked the true measurements intended to be made.

(2) The sensitive capacitor element was coupled to an oscillating circuit and the indication was obtained by detecting either an amplitude modulation or a frequency modulation in the oscillating circuit. This arrangement suffered from the disadvantage that the frequency of the oscillating circuit must be changeable or variable, and thus the frequency tended to be changed by spurious ef- Ifects, and the frequency often tended to drift. The difculty was most pronounced when an attempt was made to measure very slow changes in the capacitor element.

(3) The sensitive capacitance element formed a part of a bridge circuit in which one arm was extended electrically to a remote location and reactauce was reflected back to the bridge circuit by changes in a remotely located tuned circuit. With this arrangement the system was inherently non-linear and, to some extent was unstable, be-

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cause the system must of necessity operate on one side of a resonance curve, wherein `strictly speaking there is no truly linear slope.

In a system embodying the present invention the bridge circuit is formed by two sensitive capacitance elements and by two phantom arms and thus advantageously avoids the problems discussed above. This measurement system includes the least possible number of circuit components physically present in the bridge circuit and thus limits the sources of measurement errors. Moreover, the system rapidly responds to very slight changes in capacitance and produces an output voltage which is a faithful reproduction of the capacitance changes.

Among the many advantages of measurement systems embodying the present invention are their ability to measure minute changes in capacitance to a high degree of accuracy and their high degree of stability in operation. As a result of this marked stability there is no signicant output signal produced, except when there is a change in the balance of the bridge due to a change in the condition being observed. Moreover, the system provides a linear, accurate response over a Wide range of frequencies, in this example extending from zero cycles per second up to twenty thousand cycles per second.

Among the other advantages of measurement systems embodying the present invention are those resulting from the yfact that they are convenient to calibrate, ruggedly resisting ambient conditions, and are adapted for use in a wide variety of installations.

Other features and objects of the present invention and the many attendant advantages provided by use of the invention will be appreciated as the same becomes more fully understood lfrom a reference to the example of a measurement system described herein as embodying the present invention and illustrated in the accompanying drawings, in which:

FIGURE 1 is a functional sche-matic block diagram of a measurement system embodying the present system;

FIGURE 2 is a schematic circuit diagram of this measurement system of FIGURE 1;

FIGURE 3 is an elevational sectional view of a transducer in the form of a rectilinear accelerometer, with the bridge circuit being incorporated;

FIGURE 4 is an elevational sectional view of another transducer for measuring slight deflections of various structures under load;

FIGURE 5 is a simplified schematic circuit diagram of portions of the system of FIGURES l and 2 for purposes of illustration and explanation;

FIGURES 6A, 6B, and 6C are simplied schematic circuit diagrams of the driver circuit, bridge circuit, and signal circuits for purposes of explanation; n

FIGURES 7A and 7B are further explanatory circuit diagrams showing the results of a small differential change in the two capacitance arms of the bridge;

FIGURE 8 is an enlarged schematic circuit diagram of the phase and amplitude sensitive discriminator of FIGURES 1 and 2;

FIGURE 8A shows the capacitance loop circuit of this discriminator;

FIGURES 9A, 9B, 9C, and 9D are simplified schematic circuit diagrams of the equivalent coupled circuits for the phase and amplitude sensitive discriminator;

FIGURE l0 shows the waveform of the voltage eI of FIGURES 7A and 7B; and

FIGURE 1l is a plot showing the over-all frequency response curve for the measurement system up to the output terminal 21.

This system of the present invention has many advantages over the four-capacitance bridge circuit shown in U.S. Patent No. 2,611,021 issued September 16, 1952, as will appear from the following illustrative example. This present system enables a more convenient and rapid adjustment into initial balance, and is markedly more stable in operation. Moreover, this present system is considerably less complex while being more sensitive and linear in output.

General description of the system and the operation As shown in FIGURE 1, the measurement system of the present invention includes a bi-resonant bridge 2 having two capacitance arms and two phantom arms. That is, two arms of the bridge are not physically present within the bridge circuit but they are electrically present within the bridge circuit, being reflected into the bridge circuit from other portions of the measurement system. The bridge 2 is energized by a suitable alternating current carrier signal ofl fixed frequency. For example, in this systern the carrier signal has a low radio frequency of 456 kilocycles. This carrier signal is transmitted to the bridge circuit by means of a low-impedance transmission line schematically illustrated at 3 from a crystal-controlled oscillator and driver amplifier circuit 4.

The bridge circuit 2 is initially balanced, and resonance at the carrier frequency is induced in two modes. In response to the condition or phenomenon being measured, generally indicated at 1, the two capacitance arms are changed differentially, and an unbalance signal voltage is developed at the two output terminals of the bridge. This unbalance signal voltage is transmitted by means of another low-impedance transmission line 5 to a signal amplier 6. As indicated by a connection 7, the amplified unbalance signal voltage is fed into null indicator and meter circuits 8 which provide a visual indication to the operator of the state of balance or unbalance in the bridge circuit. The amplified unbalance signal is also fed through a pair of connections 9 and 10 from the signal amplifier to a phase and amplitude sensitive discriminator 12. In order to provide a reference signal for the discriminator 12, the carrier signal is supplied through a transmission line 11 and an output reversing switch 13 to an injection amplifier 14 and is amplified and fed through connections illustrated at 15 and 16 to the discriminator 12.

In this discriminator 12 the amplified unbalance voltage from the bridge is compared with the amplified carrier signal. By means of phase comparison, the sense or direction of the unbalance of the bridge circuit is determined. The amplitude of the unbalance voltage is sensed; the carrier signal and side-band components are removed by rectification; the current in one-half of the discriminator is subtracted from the current in the other half; and a direct-current difference voltage of the proper amplitude and polarity to indicate accurately the condition or phenomenon 1 is supplied through leads 17 and 19 to an output stage 18. From the output stage, the output voltage is then sent through a lead 21 to the desired utilization circuits 20, which may include various instruments such as recorders, controllers, meters, etc., which suitably utilize the information obtained concerning the observed condition 1. In order to indicate the amplitude of the carrier signal, there is a connection 25 from the oscillator and driver amplifier to the meter circuits 8.

Transducers incorporating bridge circuit Before considering in detail the operation and construction of the system of FIGURES 1 and 2 it may be helpful to direct attention to FIGURE 3 which shows a suitable transducer in the form of a rectilinear accelerometer with the bridge circuit of the system built into the transducer, and to FIGURE 4 which shows another transducer suitable for measuring small displacements in which the bridge circuit is also incorporaed. As shown in FIGURE 3, the rectilinear acceleration transducer 22. is sensing the acceleration of the structural member 1 of a vehicle. This transducer is mounted rigidly onto the structure 1 by a mounting stud 23 screwed into the base 24 of a housing 26 formed of rigid material, for example stainless steel. Within the housing are a rigid partition 28 and a separator 30, each of electrically conducting material, such as hard brass, forming three electrically shielded compartments 31, 32, and 33. In the upper or sensing compartment 31 is a compliant cantilever 35 of electrically conducting material, such as beryllium copper, which projects inwardly from the housing 26 and supports a mass M at its free end having a pair of capacitor plates 36 and 38 formed on its opposite sides. Adjacent to either side of the mass M are two capacitor plates 40 and 42, spaced uniformly from the plates 36 and 38, respectively, and mounted upon rigid insulation supports 43 and. 44, thus forming a pair of differential air-gap capacitors 45 and 46.

In order to energize the capacitors 45 and 46, a lowimpedance transmission line 3, shown as a shielded link coupling, is connected to the primary winding 47 of a bridge transformer 48 having a center-tapped secondary winding 49. A pair of leads 50 and 52 extend from opposite sides of the secondary 49 through small openings in the partition 2S to the two fixed plates 40 and 42, respectively. An alternating voltage eb is applied across the leads 5t) and 52 and forms the bridge-driving voltage. The bridge transformer 48 is constructed with an adjustable powdered iron core as indicated by the adjustable symbol and enables a fine adjustment to be made to assure that the center tap has a voltage which is exactly centered between the signal on the leads 50 and 52. As a result of the differential change in the capacitance of the capacitors 45 and 46, this system produces an output voltage in the lead 21 to the utilization circuits 2t) which accurately and linearly indicates the magnitude and direction of the acceleration.

When acceleration occurs in an axial plane of the mounting stud, then the mass M moves closer to one of the fixed plates 40 or 42 and farther from the other, causing a differential change in the capacitances of the air-gap capacitors 45 and 46. In order to sense the slight changes in the capacitances, a signal transformer 54 has its primary winding 55 connected by a lead 56 to the point b at the fixed end of the cantilever 35 and by a lead 57 to the point a at the center tap of the secondary 49. The secondary 58 of the signal transformer is connected to a low impedance transmission line 5, illustrated as a shielded link coupling. For obtaining to the fullest the advantages that can be provided by this system, it is important that good shielding be maintained between the driver and signal transformers 4S and 54. This shielding should prevent both stray electromagnetic as well as stray electrostatic coupling, and is provided by the relatively thick conductive partition 28 and divider 3f), which are quite effective in blocking the low radio frequency signal. At lower frequencies magnetically permeable shields may be used in addition to the conductive shields.

It will be noted that the transducer 22 incorporates the bridge circuit 2, which includes the two capacitance arms 45 and 46 and also the two phantom arms 59 and 6i) (please see also FIGURE 2). The fixed end of the cantilever 35 is secured to the housing 26, and this constitutes a common return (ground) connection for the lead 56. In order to assure that the various elements remain in fixed positions within the three compartments 31, 32, and 33, they are filled after adjustment of the transformers 48 and 54 with a rigidly setting insulating plastic material or potting compound, for example an epoxy resin.

The displacement-sensing transducer 22A of FIGURE 4 is generally quite similar to the transducer 22, and parts performing corresponding functions have corresponding reference numbers. In operation this transducer is used to measure very small deflections of structural members under load, for example, to measure the deflection of steel beam 1 under stress. The base 24 is anchored by the mounting stud 23 to a fixed support 62. To sense the deflection of the structure 1, such as a pavement strip, a rigid cantilever 64 of insulation material projects from the structure' 1 through a window 65 in the housing 26 and connects to the end of a stiff conductive cantilever 66 which includes the capacitive electrodes 36 and 38. The lead 56 is connected to the outer end of the conductive cantilever 66, and a flexible ground connection extends from this cantilever to the housing 26. As the member 1 defiects very slightly under heavy load, the movement causes the capacitance of one of the airgap capacitors 45 or 46 to increase while the other decreases. As a result, the utilization circuit is supplied a voltage which corresponds with the magnitude and direction of the deiiection.

Oscillator and driver amplifier A system which has proven to be very sensitive and accurate in operation is shown in FIGUREl 2. The oscillator and driver amplifier circuit is generally indicated within the dashed rectangle 4 and includes an oscillator stage 67 utilizing a dual triocle 68 having both sections of the tube coupled through a common cathode resistor 69. A crystal 70 which is resonant at 456 lrilocycles is connected to the common return (ground) circuit 71 and is connected through a resistor 72 and across a gridreturn resistor 73 to the grid of one section of the dual triode. The anode is supplied from a suitable voltage supply source B+ through a filter resistor 74 and is effectively tied to the ground 71 through a filter capacitor 76. This oscillator 67 is the so-called Boot Strap oscillator and is characterized by an extremely stable output frequency. This oscillator was described in detail by applicant in the January 1953 issue of Electronics, but its features are described briefly here for convenience. The crystal 70 serves a dual purpose. It serves as an extremely sharp filter, thus providing excellent waveform at the output coupling capacitor 83. Also, it is the frequency-determining circuit element for the oscillator. The oscillator is self-starting when the anode supply voltage is turned on.

There is no phase reversal in the two sections of the dual triode. The voltage across the crystal is fed to the grid 75 of the right triode section, which operates as a cathode-follower and drives the left triode section by means of the coupling through the common cathode resistor 69. The voltage of the anode 79 thus changes in the same direction as the grid 75, developing a signal across the plate load including the inductor 77 shunted by the capacitor 78. This plate signal is fed back to the crystal 70 and to the grid 75 through a positive feedback circuit including a coupling capacitor 80 and a resistor 31.

The purpose of the inductor 77 and capacitor 78 is to offset the distributed plate capacitance and transit time of the left triode section so as to make the positive feedback voltage directly in phase with the voltage across the crystal 70. In this example it is found to be advantageous to have the resonant frequency of the parallel circuit including the inductor 77 and capacitor 78 detuned just slightly to offset the transit time, being resonant at a frequency slightly below the oscillator frequency. Thus, the plate circuit 77, 78 provides a leading phase angle which advances the phase of the voltage of the anode 79 and compensates for any delay in the left triode due to transit time and distributed plate capacitance.

lIt will be understood that the voltage supply source B+ is a typical power supply, which in this example, supplies a positive direct current voltage at the terminals B+ of 300 volts with respect to the common return or ground connection 71. This power supply also has a terminal B+, which is seen at the lower right in FIGURE 2 and supplies a negative D.C. voltage of minus 150 volts with respect to the common ground connection.

From this oscillator stage 67 the carrier signal is coupled through a capacitor 83 and a resistor 84 to a potentiometer 85 which is used for adjusting the amplitude, i.e. level, of the carrier signal which is then amplified by the tuned push-pull amplifier 86. In the input of this amplifier is a triode 38 with its grid connected to the adjustable contact 89 of the potentiometer and having a cathode-bias resistor 90 with a bypass capacitor 91. Anode voltage is supplied to this triode through a lead 92 and a tuned plate load including an adjustable capacitor 93 and an inductor 94. The tube 88 drives a tuned pushpull grid circuit 96 which acts as an accurate phase splitter yand avoids reflecting any Miller effect from the triode 88 into the oscillator stage.

From the anode the carrier signal is coupled through a capacitor 97 to a tuned push-pull grid circuit including an inductor 98 having -a center tap connected through a resistor 99 and a lead 100 to the common ground. A pair of fixed capacitors 101 and 102 shunted by small adjustable -capacitors 103 and 104, respectively, are connected from the lead to a pair of grid leads 105 and 106 which are connected to opposite sides of the inductor 98 and control the push-pull pentodes 107 and 108. A capacitor 109 is shunted between these leads 105 and 106. Anode potential for these tubes is supplied from B+ through a filter resistor 111 and over a filter capacitor 112 to a center tap in the primary winding 114 of a driver transformer 115 having an adjustable powdered iron core. The opposite sides of the primary are connected through leads 117 and 118 to the respective anodes of the tubes 107 and 108. A resistor 119 supplies screengrid potential from the center tap to the screen grids, and a common cathode resistor 120 provides the operating bias for the grids. For providing an indication of carrier level, a resistor 121 and capacitor 122 are connected in series from the anode lead 118 to the connection 25 which runs finto the meter circuits 8, as will be described in detail further below.

To energize the bridge circuit 2, the secondary winding 124 of the driver transformer 115 is connected by a shielded low impedance transmission line 3 to the primary 47 of the bridge transformer 48. It is preferred, for the most sensitive and accurate response, that an exact electrical symmetry be established in the opposite halves of the primary of the driver transformer 115 and in the opposite halves of the secondary of the bridge transformer 48. Also, the primary winding 114 and the secondary winding 49 are resonant `at the frequency of operation with the capacitance associated therewith, as will be explained. The primary winding 114 is -resonant with a capacitance C1 thereacross, which includes the stray plate capacitance of the tubes. Although the driver amplifier is shown as being push-pull, for the practical reason that this reduces distortion, the driver amplifier may be single-ended, and the primary 114 may be unsymmetrical; however, the system as shown provides very sensitive precise measurements.

Further description of the system and features of operati-0n In order to explain the operation of the system which is shown in FIGURES 1 and 2, attention is directed to the schematic circuit diagram of FGURE 5, which illustrates in simplified form portions of this system. The push-pull pentodes 107 Iand 108 develop a driving voltage ed between the leads 117 and 118 and across the primary winding 114 which has a self-inductance L1. The capacitance C1 and the inductance L1 are resonant at the frequency of operation. Also, there is a mutual inductance M1 between the primary 114 and the secondary 124.

The bridge transformer 48 is identical to the driver transformer 115, except that the respective positions of the primary and secondary are reversed. Its secondary has a self-inductance of L1 and there is a mutual inductance of M1 between its primary and secondary. In operation, the two capacitors 45 and 46 `are equal and are in series across the secondary winding 49 and are resonant with the inductance L1 of the winding 49. Thus, each of these capacitors normally has a capacitance Value of 2C1.

As discussed above, the bridge transformer is adjusted so that the point a is an exact electrical center point with respect to the leads 50 and 52, and normally the two capacitors 45 and 46 are equal. Thus, regardless of the magnitude of the bridge voltage eb, appearing between the points m and n, the signal voltage es appearing between the points a and [1, when the bridge is balanced, is zero. As a consequence, the resonance of the secondary 49 with the two capacitors 4S and 46 is not in any way affected by the presence or absence of the connections 56 and 57 to the primary 55 of the signal transformer. Thus, the operation of the -bridge circuit 2 .is extremely stable, because it is not significantly influenced by the operation of the signal output circuit.

FIGURE 2 shows the actual physical arrangement of the leads 50 and 52 which are connected from opposite sides of the secondary 49 to the opposite corners m and n of the bridge. However, as indicated by the dashed lines in FIGURE 2, the bridge circuit operatively includes the impedances 59 and 60 each shown by the symbol Z, which are reflected into the bridge circuit from the primary side of the bridge exciting transformer 48. Consequently, as is illustrated in FIGURE 5, the bridge circuit operates as if the phantom arms 59 and 60 were connected between the center tap of the secondary 49 and the bridge points m and n.

As seen in FIGURE 5, the two capacitors 45 and 46 are effectively connected by the respective phantom arms 59 and 69 in parallel with one another between the points a and 1). Under these conditions, with the capacitors 4S and 46 resonant with the secondary 49 and with no voltage between the points a and b, then the signal circuit can be effectively resonant with the capacitors 45 and 46 even though there is no voltage developed across it and no current in it. To provide this resonance, the primary 55 of the signal transformer has an inductance value L2 which will resonate with a capacitance value of 4C1, which is formed by 2C, in parallel with 2C1. In other words, L2 is one-quarter of L1 iin value.

As shown in FIGURE 5, the secondary of the signal transformer is connected to the primary 126 of a receiver transformer 128, which is identical with the signal transformer except that the relative positions of primary and secondary are reversed. The secondary 129 of the receiver transformer has the inductance value of L2, which resonates at the carrier frequency with a parallel capacitor 130 of value C2, equal to 4C1. In the signal amplier 6, the output voltage er from the receiver transformer is amplified and fed as a balanced push-pull signal into the X1 and X2 terminals of the phase and amplitude sensitive discriminator 12.

To supply a reference signal :to this discriminator, the carrier signal is fed from the secondary of the driver transformer over a low impedance line 11. This reference signal passes through a reversing switch 13 into a high resistance potentiometer 131 which is across the primary 132 of a transformer 133. Its secondary 134 forms the input of a balanced injection amplifier 14, which provides a balanced push-pull carrier signal to the Y1 and Y2 terminals of the discriminator 12. The operating advantages of the phase and amplitude sensitive discriminator are explained in detail further below.

Referring to the schematic circuit diagram of FIGURE 6A, the driver amplifier tubes 107 and 108 are most easily schematically illustrated by an equivalent singleended amplifier, this being shown symbolically as a generator with zero internal impedance delivering a signal voltage e and having a resistor rp in series therewith, representing the effective dynamic plate resistance of the driver amplifier. It will be understood that in FIGURE 6A for convenience of illustration the link coupling line 11 and injection amplifier connected to the line 3 are only briefly indicated.

In order to obtain best results, the effective dynamic plate resistance rp of this amplifier should be quite high, preferably at least l megohm. The reason for this is that the Q for the circuits should not go much below 8 or S 10 in order to obtain the best response, and this Q is determined by the value of rp. There is no particular advantage in seeking a Q value above 20, and a Q value of 4 or lower causes poor operation. The impedances Z at 59 and 60 comprising the phantom arms each include the resistances and re.

T he resistor r1 represents the dissipation resistance that is inherent in the primary 114, and the two resistors represent the dissipation resistance of .the two halves of the secondary winding 49. The two resistors are the effective resistances included in the impedances Z of the two phantom arms 59 and 60, resulting from the dynamic plate resistance rp of the amplifier reflected into the bridge.

FIGURE 6B is a schematic diagram of a circuit arrangement which is precisely equivalent to the circuit of FIGURE 6A, and wherein the capacitor C1 is in series with an equivalent signal voltage e. In order to show that this -is true, it is noted that if the circuit of FIGURE 6A is cut open at the two points F and G, the opencircuit voltage e0 presented is:

If the circuit of FIGURE 6B were cut open at the points F and G', the equivalent open-circuit voltage e would appear.

The open-circuit impedance looking toward the left from points F and G in FIGURE 6A is:

This should be equal to the open-circuit impedance looking toward the left from the points F and G in FIGURE 6B, which is:

In view of the fact that the second term in the denominator of the right expression is so very small compared with the first term, this can be simplified, as stated, to:

1 (5) Te- @20x27-D The circuit of FIGURE 6B can be further simplified to another precisely equivalent form, assuming that the losses in the link coupling circuit are very small, which is the case here. This conversion for link-coupled circuits is shown in Radio Engineers Handbook by F. E. Terman, First Edition, Fourth Impression (1943) on page 16'2, and is shown in FIGURE 6C.

In addition to the electrical symmetry shown in FIG- URES 6A, 6B, and 6C, it is important to have a coupling coefficient k which is exactly, or very nearly, equal to the critical coefiicient of coupling kc. This produces the maximum lpossible voltage eb across the capacitors 45 and 46, which voltage is related to the voltage e' in series 9 with the primary by the following expression, derived from page 156 of the Radio Engineers Handbook:

61, Q l (6) e- 2 2lic Also, this critical coupling produces the desirable broadened fiat-topped curve which is characteristic of the response in the secondary circuit of critically coupled resonant circuits. Thus, the system is not sensitive to changes in the frequency of the carrier signal over a range of 4%.

In order to explain the operation, assume that the observed condition 1 now causes a small differential change 6c to occur in the two air-gap capacitors, so that one of them becomes 2C1-i-c and the other becomes 2CV-5c. A voltage difference es now develops across the :mints a and 17.

In this example, c is the instantaneous value of the incremental change in capacitance which is assumed to vary sinusoidally with time. The value of the small voltage difference es, as appears from FIGURE 6C, is:

c (7) es: @bm

and this signal es is applied to the primary 55 of the signal transformer 54, as shown in FIGURES 2 and 5 connected to points a and b,

In the circuit diagram of FIGURE 7A, this voltage es is shown as being delivered from a generator having zero internal impedance connected in series with the two sides of the bridge 2, which are in parallel.

FIGURE 7B shows a symmetrical double-tuned circuit which is the electrical equivalent of that shown in FIG- URE 7A, assuming that the losses in the link coupling 5 are small. This conversion is analogous to that made in going from FIGURE 6B to 6C. The requirement for electrical symmetry exists for the signal output transformer 54- and for the receiver transformer 128 and the link coupling 5 therebetween, as was true of the bridge input circuit shown in FIGURE 6A. Also, for optimum performance, it is preferred that the coefiicient of coupling k be equal to the critical coefficient kc.

Then, the voltage er in FIGURE 7B across the secondary of the receiver transformer is derived from Terman, page 156, as above, and is:

where A is a linear coefficient of amplitude.

Substituting from Equation 7, this expression becomes:

ceiver transformer 128 is thus seen to have the following value, as derived from Equations 9 and l0:

Anno se1 czAC cos wmf Eb cos @et cos wmt=Am cos met cos wmf where Eb is the maximum value of the bridge driving voltage, cos wat is the angular frequency of the carrier or bridge driving voltage, and Am is a linear coefficient of amplitude for the modulated signal.

This receiver voltage er is amplified by the signal amplifier 6 and is impressed as a balanced push-pull voltage upon the terminals X1 and X2 of the phase and amplitude sensitive discriminator circuit 12.

Under the conditions described, the receiver voltage er has the form shown in FIGURE l0. It is noted that voltage of the carrier frequency and of the modulating frequency are completely absent, and there are two equiamplitude voltages of side-band frequencies present. These side-band frequencies are symmetrically spaced on a linear frequency spectrum by an amount wm on either side of the angular carrier frequency we. This can also be seen by manipulation of Equation 11 in accordance with the trigonometric relationship:

Accordingly, Equation 11 can be rewritten:

e :Ao/10E,

As a result of these advantageous relationships, the signal amplifier 6 is required to amplify only a relatively narrow band of frequencies, namely twice the over-all frequency response, centered about the carrier frequency of 456 kc.

Detailed description of signal amplier circuit 6 As seen in FIGURE 2, the signal amplifier 6 includes a pentode in its first stage having a plate supply decoupling resistor 136 and filter capacitor 138, with a cathode resistor 139. The screen voltage is provided by the dropping resistors 149 and 141 and the filter capacitor 142. The amplified signal voltage appearing across the anode resistor 143 is fed through a coupling capacitor 144 to a potentiometer 145 having its adjustable Contact 146 connected through a grid resistor 147 to the grid of a pentode 148 in the second stage. This second stage is identical to the first and circuit components performing functions corresponding to the first stage have corresponding reference numbers.

From the output coupling capacitor 144 of the second stage the amplified signal is applied across a resistor 149 to the grid of a pentode 151i in a tuned push-pull amplifier including a companion pentode 151. The two sides of this push-pull amplifier are identical except for a minor difference which improves the balance of the output voltages on the two leads 9 and 19 to the phase and amplitude sensitive discriminator and demodulator 12. There is a capacitor 153 shunting the resistor 149 which is connected to the grid of the other pentode 151. Each tube has a cathode bias resistor 163 shunted by a by-pass capacitor 154 and connected to the common return circuit through a parallel tuned circuit 155 and 156, which is resonant to the carrier frequency. The anode voltage is supplied from the source B+ through a series resistor 157 and across a filter capacitor 158 to the center tap of the primary 159 of the output transformer 160. The screen grid voltage is supplied from B+ through a filter resistor 161 and past a filter capacitor 162 and through a parallel resonant circuit 152 which is tuned to the carrier frequency. The anodes of the tubes 156 and 151 are connected to opposite sides of the primary 159 which has a self inductance of L3 and is resonant at the carrier frequency with a capacitor 191B having a capacitance value of C3. From the opposite sides of the secondary 191, having a grounded centertap 192, a balanced push-pull signal voltage is fed through the leads 9 and 16 to the X1 and X2 terminals of the phase and amplitude sensitive discriminator 12.

Injection amplifier circuit 14 As seen in FIGURE 2, the injection amplifier circuit 14 is generally similar to the final stage of the signal amplifier circuit. The secondary 134 is connected across an adjustable phase balance circuit comprising a fixed capacitor 164 shunted by an adjustable capacitor 165. One side of these capacitors is connected to the grid of a pentode 166 in a tuned push-pull amplification stage including a companion pentode 168 which has its grid connected to the opposite side of the capacitors. Each tube 166 and 168 has a cathode bias resistor 170 shunted by a capacitor 171 and connected to the return circuit 71 through a tuned parallel circuit which is resonant at the carrier frequency and comprises an inductor 172 and a capacitor 173. The anodes are energized from B+ through a resistor 174 connected to a filter capacitor 175 and connected to a center tap on the primary 176 of the output transformer 177. The screen grid energizing circuit has a resistor 178 and a filter capacitor 179 connected to a tuned circuit 180 which is resonant at the carrier frequency. A capacitor 181 is connected across the primary 176 and is resonant therewith at the carrier frequency. The amplified carrier signal is fed in balanced push-pull relationship from the secondary 182 through the leads 15 and 16 to the respective Y1 and Y2 terminals of the discriminator 12. This secondary has a grounded center connection 193.

Features of operation of the phase and amplitude sensitive discriminator circuit J2 In order to explain the operation of the phase and amplitude discriminator circuit 12, it is convenient to consider the circuit 12 as including three interconnected networks, as illustrated in FIGURES 2, 8, and 8A. There is a resonant capacitance network loop 200 comprising eight identical capacitors 201 through 208, which includes the pair of signal input terminals X1 and X2 and the pair 0f reference input terminals Y1 and Y2. These eight capacitors are in series with one another around the loop, as seen in FIGURE 2, and there are two of them in series between each of the four input terminals in succession around the loop. In FIGURE 8A, this capacitance loop 200 is redrawn separately and arranged with the X1 and X2 terminals at the top and the Y1 and Y2 terminals at the bottom, which is convenient for purposes of illustrating the operation.

The second network, within the phase and amplitude sensitive discriminator 12, as seen most clearly in FIG- URE 8, is a rectifier network or diode matrix 210 including eight rectifiers 211 through 218, inclusive, and co-operating with the capacitance network to form four half-wave voltage-doubling circuits. These rectifiers are advantageously arranged to provide a reference each half cycle to the reference point G and the lead 17, which is indicated as the D.C. Reference lead. The rectifier network 210 has four input terminals A, B, C, and D respectively which are connected to the capacitance loop 200 at the points A, B, C, and D, and there are two output terminals E and F from the rectifier network.

The third network within the circuit 12 is a four-legged resistance bridge 220 including the identical resistors 221, 222, 223 and 224. This resistance bridge effectively compares the D.C. Voltage at point E with the D.C. voltage at point F and provides an output voltage on the terminal lead 19 which is proportional to the difference between the voltages at E and F. A capacitor 225 extends diagonally across the bridge and provides smoothing for the output voltage on the lead 19.

Before considering further aspects of the operation of the discriminator c rcuit 12, attention is directed to FIGURES 9A and 9B, which are schematic circuit diagrams illustrating the desirable resonance between the output stage of the signal amplifier 6 and the capacitor loop 200. In FIGURE 9A, the signal amplifier tubes 146 and 148 are represented as an equivalent single-sided amplifier for reasons of simplifying the illustration, and this singlesided amplifier is shown symbolically as a generator with zero internal impedance delivering a voltage ea and having a resistor rpa in series therewith as caused by the dynamic plate resistance of these tubes. The capacitance C2 represents the effect of the plate capacitance of the tubes 146 and 148 and the capacitor 190 in the anode circuit, and r3 represents the dissipation in the anode circuits. The self-inductance of the primary 159 has a value of L3, resonant with C2, and the secondary 191 also has a selfinductance value of L3, with a mutual inductance of M3 therebetween. The two resistors represent the dissipation in the two halves of the secondary, and the two resistors are the effective resistances coupled into the secondary caused by the dynamic plate resistance rpa. The eight capacitors 201-208 each has a capacitance value of 2C3.

FIGURE 9B is precisely equivalent to FIGURE 9A. It will be appreciated that the eight capacitances of value 2C3 in the capacitance network 200 are effectively in a series parallel relationship between the input terminals X1 and X2. Thus, advantageously, the capacitance loop 200 provides an overall effective capacitance value of C2 between the points X1 and X2, which is the proper value to resonate with the output stage of the signal amplifier.

It is convenient to consider the signal voltage applied across the terminals X1 and X2 as having a value of Zex, as is also indicated in FIGURE 8A.

Similar considerations to these apply to the schematic circuit diagrams of FIGURES 9C and 9D which represent the interaction of the output of the injection amplifier 14 and the circuit 12. Consequently, the capacitor loop 200 is also resonant with the output stage of the injection amplifier 14. As a result of these advantageous interactions, the discriminator circuit 12 requires very little energization compared with the magnitude of the output obtained. The only energization required from the amplifier circuits 6 and 14 is that necessary to maintain oscillations in the capacitor loop 200. It is also convenient to consider the injected carrier signal Voltage across the terminals Y1 and Y2 as having a value Zey, as is also indicated in FIGURE 8A.

In operation, as shown in FIGURE 8A, the signal amplifier output voltage 26X acts between the terminals X1 and X2. During one-half cycle the terminal X1 is at a voltage of -f-ex, and the other at -ex. The four capacitors in series 201, 202, 203, and 204 act like a voltage divider so that the points A and B tend to he at and 205 form a second voltage divider so that the points C and D tend to be at ieg 2 and respectively. With respect to the X1 and X2 terminals, both the Y1 and Y2 terminals tend to be balance points. This relationship is also seen clearly from FIGURE 8, wherein the capacitance loop is opened at the X1 and X2 terminals to provide the convenient configuration as illustrated therein.

Additionally, with an injection reference voltage of -l-ey and C and D each tend to be at a voltage of Accordingly, the signal and carrier voltages are additive at points A and D and are in opposition at the points B and C. When the relative phase of the signal voltage is reversed with respect to the carrier signal, then the respective effects are reversed.

When the terminal X1 is positive with respect to X2, then current flows from X1 (upper left in FIGURE 8) down through the capacitor 201 as displacement current to point A, through the diode rectifier 211 to point E, and down through the resistor 221 to the reference point G which is connected to the lead 17. Current flows from the point G down through the resistor 224 to the point F and through the diode rectier 216 to the point B and as displacement current through the capacitor 204 to the terminal X2 (lower left). Thus, during this half cycle when X1 is positive and X2 is negative, the point E tends to become positive with respect to F.

Assuming that the system has been in operation for several cycles of the carrier signal, then the rectification by the diode rectiers will have built up charges on the capacitors as shown. The Way in which these charges are obtained will be explained further on, but it is noted that these charges provide a desirable voltage-doubling effect.

During this half-cycle of the signal voltage when the terminal X1 is positive with respect to X2, attention is also directed to the lower right terminal X1 in FIGURE 8. It will be noted that current flows upwardly through the capacitor 268 to the point C, which is effectively tied to the reference point G by the rectifier 214. Thus, the capacitor S is receiving a charge with polarity as shown. Similarly, the point D is effectively tied to the reference point G by the rectifier 213, and so the capacitor 205 also receives a charge, of polarity as shown.

When the second half-cycle of the signal voltage is occurring, the applied voltages are all reversed from those shown in FIGURE 8. Then the charges which were established on the capacitors 268 and 205 act in series aiding relationship with the applied voltage so as to provide a voltage doubling effect. Current ilows from the upper right terminal X2 (now positive) as a displacement current down through the capacitor 265, through the rectifier 212, and down through the resistor 221 from point E to the reference point G. From G, current tiows down through the resistor 224 and continues on through the rectifier 215 and the capacitor 293 to the terminal X1 (lower right). Thus, the point E remains positive with respect to the point F.

During this second half-cycle, the points A and B are effectively tied to the reference point G by the rectifiers 21S and 217, respectively, and so the capacitors 2&1 and 264 receive a charge, as was indicated above.

There is no output signal on the lead 19 in the absence of the carrier reference voltage, for the bridge 2213 has four equal resistance arms, and the positive voltage at E exactly offsets the negative voltage at F.

From the above discussion, it will be understood that when the 4carrier injection voltage is applied With phase relationships such that the terminal Y1 is positive with respect to Y2, then the capacitors 262, 263, 297, and 2% become charged as shown and provide desirable voltage doubling action. When Y1 is positive, then B and D are tied to the reference point G by the rectiiiers 217 and 213, respectively. During the subsequent half cycle, the points A and C are referenced to G through the rectifiers 218 and 214, respectively.

lli

When the relative phase of the signal voltage eX has the relationship to the carrier voltage ey as shown, then the signal and carrier voltages are additive at the points A and D and oppose each other at the points B and C. As a result, the magnitude yof the positive voltage at the terminal E becomes larger than the magnitude of the negative voltage at the point F. The output sensing bridge 220 becomes unbalanced and provides a positive D.C. output voltage on the lead 19 Which is proportional to the extent of the unbalance of the condition-sensing bridge 2.

When the capacitance bridge 2 is unbalanced in the opposite direction, then the phase of the signal voltage reverses with respect to the carrier voltage, and the D.C. output signal on the lead 19 becomes negative and proportional to the unbalance of the capacitance bridge 2.

It is desired to direct attention to several advantages of this system. The modulation of the carrier by the biresonant capacitance bridge 2 is accomplished by the addition of the carrier and two symmetrical sideband components. The carrier is suppressed when the bridge initially balanced so that the resultant output includes only the two symmetrical sideband components, as shown in FIGURE l0 and in Equation 13. A voltage of the angular modulating frequency wm is not used in the process and conventional tuned circuits and other nonlinear circuit elements are avoided.

In the discriminator at the respective points A, B, C, and D, four modulated carrier voltages are present and also four injected carrier voltages are present. The rectification of these four modulated carrier Waves which are paired in push-pull (contra-phase) relationship, results in direct cancellation of even-harmonic distortion of the component of angular modulating frequency om. The result of this action is the accomplishment of dist-ortionless operation and linear response, insofar as the frequency of the condition l under observation is concerned.

Output stage and utilization circuits The direct current output signal on the lead 19 is `fed to the grid of the left section of a dual triode 236, with its cathode connected through a resistor 232 to a micro-ammeter 234. This is a center-zero type meter, and its other terminal is connected to the cathode of the right triode section. Both cathodes are connected through identical resistors 235 and 236, respectively, to the B- voltage source.

In order to provide a D.C. reference voltage source, a voltage dropping resistor 23S is connected from the B- source to a pair of identical potentiometers 239 and 240, and the other sides of these potentiometers are connected to the common return or ground circuit. The lead 17 is connected to the adjustable Contact of the potentiometer 239, for adjusting the D.C. reference vol*- age, and a by-pass capacitor 242 is connected from lead 17 to ground.

For adjusting the position of the pointer of the meter 234 in the absence of any output voltage, the grid of the second triode section is connected to the contact of the potentiometer 240.

A suitable utilization circuit, such as a measurement recording and control device 2t) is connected by the lead 21 to the cathode of the iirst triode section.

N all indicator and meter circuits In order to Ifacilitate the balancing of the sensing bridge 2 and to show the carrier level, the null indicator and meter circuits 8 are provided. The amplitude `of the carrier is indicated by a meter 259, as shown in FIGURE 2. The carrier signal is fed by the lead 25 to the grid of a triode amplifier stage 252, which includes a grid return resistor 253, a cathode resistor 254, by-pass capacitor 255, and an anode load resistor 256. The negativegoing excursions of the carrier signal appearing at the anode of the tube 252 are rectified by a diode 257, creating a charge of the polarity as shown on a capacitor 258. The gas tube 259 is included to protect the diodes 257 and 260 against transient surges. The positive-going excursions of the amplified carrier signal are fed through a rectier 260 and a series resistor 261 to the meter 250.

For indicating the null condition of the bridge 2, a reflex double amplifier circuit 262 is used. That is, the voltage gain of a pentode 264 is used at the carrier frequency, and at the same time it serves as a D.-C. amplifier for the metering circuit including the meter 265. This meter circuit is designed so that the meter scale is approximately logarithmic on an inverse scale.

The amplified signal from the sensing bridge 2 is fed from an intermediate point in the signal amplifier 6 (please see FIGURE 2) through a resistor 269 and an OFF-READ switch 270 and through the lead 7 and a coupling capacitor 271 to the grid of the pentode 264.

The parallel resonant circuit 271 and 272 enhances the gain of the pentode at carrier frequency and reduces the A.-C. gain at sub-harmonic and upper-harmonic frequencies. This same resonant circuit serves to block the amplified carrier current from the metering circuit. The cathode follower stage 274 is employed as a low-impedance reference point for the positive terminal of the meter 265. The negative terminal of this meter connects to the anode of the pentode 264 through the low D.-C. resistance path provided by the inductor 271. Any carrier current which happens to pass through the tuned circuit 271', 272 is by-passed to ground by the capacitor 276.

The metering circuit operates as a low-resistance voltmeter that indicates the average D.C. anode potential of the pentode 264 referred to the potential level of the cathode of the cathode-follower stage 274. The left section of the duo-diode 278 is shunted by a resistor 277 and is coupled to the anode of the pentode 264 by a capacitor 279. This left diode section serves as a halfwave rectifier for carrier frequency, imposing the average D.-C. component of the rectified carrier voltage on the grid of the pentode via the resistances 281 and 282 that constitute the grid return path. Ripple voltages are bypassed to ground by a capacitor 280. Thus, the pentode acts to reduce its own gain as the signal increases, and

vice versa.

The right section of the duo-diode 278 acts as an overload shunt for the meter 265 for large signal voltages. When the bridge is balanced and no carrier voltage is imposed on the pentode grid by the signal amplier, the current in the right diode section that shunts the meter is cut off by reversal of the voltage across it and the meter reads full scale, with no effective shunt. The sensitivity is very high and bridge balance may be accomplished to a high degree of precision. When the bridge is unbalanced and large voltages appear at the input to the null indicator, the current in the pentode is reduced and the voltage across the diode and meter tends to reverse polarity; and in so doing approaches zero. The conductivity and resultant low resistance of the diode is now in parallel with the meter and its series resistor. The sensitivity of the meter is greatly reduced since the diode is conducting a large share of the available current.

Simultaneously, the mutual conductance of the pentode is reduced because of the increased biasing voltage developed on its control grid by the rectified carrier voltage. As a result the anode current approaches zero. The current in the pentode cannot be cut off entirely, however, because the biasing voltage tending to cut it off is obtained from the rectified signal at its own anode. The combination of the reduction in mutual conductance and gain of the tube, and the shunting of the meter by the diode, provides a convenient near-logarithmic characteristic. When the bridge is unbalanced a large amount, any improvement in balance is readily seen; and at the same tim@ .the 4Circuit does not overload or saturate.

The adjustable potentiometer 284 provides a convenient means for adjusting the average biasing voltage and the resultant average anode potential level of the pentode. This is done so that with no signal voltage to the nullindicator circuit, the meter reads exactly full scale.

Capacitance bridge balancing method and circuits For purposes of balancing the biresonant capacitance sensing bridge 2, the capacitor 45 is shunted by a very small adjustable capacitor 290 in series with a very small fixed capacitor 291 connected in circuit between the bridge terminals b and m, with the adjustable capacitor 290 being adjacent to the grounded point b. Similarly an adjustable capacitor 294 in series with a fixed capacitor 295 is connected across the bridge capacitor 46 between the terminals b and n. Because of the small capacitance of each pair of these capacitors in series, a very small effective change in capacitance as desired can be obtained by adjustment for precisely balancing the bridge.

Remote balancing method and circuits The capacitance bridge 2 may be balanced at some physical distance by means of a cable 308, as shown in FIGURE 2 extending to a remote balancing circuit 310. A capacitor 312 is connected in series with the cable 308 at the bridge, and at the other end of the cable an adjustable resistor 313 is connected in series with an adjustable capacitor 314 and a decade capacitance unit as indicated by the capacitances at 315. By adjustment of first the adjustable resistance and then the variable capacitors 314 and 315 the bridge 2 is balanced. The resistance 313 balances the effective dissipation currents, and the capacitances 314 and 315 balance the capacitive arms of the bridge.

Calibration method and circuit In order to calibrate the bi-resonant capacitance sensing bridge 2, the calibration circuit 300 is provided shunting the capacitance arm 45 of the bridge 2. This calibration circuit includes a first relatively larger capacitor 304 shunted by a switch 302 in series with a second relatively smaller capacitor 305. The capacitance of the second capacitor 305 is relatively very small, and thus opening and closing the switch 302 causes a very small capacitance change between the points m and b.

When the switch 302 is open, the total capacitance between the points m and b includes the capacitor 304 (C4) and the capacitor 305 (C5) in series, which provide the effective capacitance:

C4 in series With O5=04C5 In this example C4 is 5,60() micro micro farads and C5 is 10 micro micro farads. Thus, the change in caapcitance AC is slightly less than 0.02 micro micro farads, which is a very small change suitable for Calibrating this sensitive circuit.

Consequently, the sensitivity of the entire measurement system can conveniently be spot-checked at any time by simply opening the Calibrating switch 302. The stray capacitance of the switch arm to ground is negligible by comparison with the large capacitance of C4 and thus does not introduce any significant error. Any excursion of the bridge from balance which is produced by the measurement of an unknown quantity may be directly compared with the excursion produced by opening and closing of the calibration switch.

In many cases it may be desired to have the calibration circuit 300 remote from the bridge 2. Then a high quality coaxial cable 306 is used of any convenient length, and

1 7 the capacitance C4 actually includes the capacitance of the capacitor 304 plus the shunt cable capacitance. The reason for the use of a high quality cable 306 is to maintain the shunt cable capacitance constant.

Example of suitable components An illustrative example of suitable components for this system is set forth below.

Capacitance (in micro Capacitors: micro farads unless noted) 45, 46 Approx. 65. 78 130.

201-208 260. 279 .001 mf. 290, 294 1-11. 291, 295 1. 258 .01 mf. 242, 271, 304 ,0056 mf 305, 312 10. 314 -140. 315 Two decades .001 mf. to .011 mf.

Inductance (in milli- Inductors: henries unless noted) 77, 271 1.175 155, 157, 161, 172, 174, 178 1 Resistance (in Resistors: megohms unless noted) 69 2.2K 72 .1 73 .39 74 4.7K S1 .22 84 .27 85 5K 90 1.5K 99, 221-224 .47 120, 139 ohms 200 111 8.2K 119 20K 121 .12 136, 145, 239, 240 K 140 .1 141 .15 147, 254, 313 1K 152 ohms 220 153, 170 do 560 157, 174 6.8K 161, 178 39K 235, 236, 233, 256 27K 261 22K 277, 281, 282 1 284 ohrns 500 From the foregoing it will be understood that the measurement system described above as an illustrative embodiment of the present invention is well suited to provide the many advantages set forth, and since many possible embodiments may be made of the various features of this invention and as the measurement system herein described may be varied in various parts, all Without departing from the scope of the invention, it is to be understood that all matter hereinbefore set forth or shown in the accompanying drawings is to be interpreted as illustrative and not in a limiting sense and that in certain instances, some of the features of the invention may be used without a corresponding use of other features, all without departing from the scope of the invention.

What is claimed is:

1. A measurement system including a biresonant capacitance bridge and a resonant phase and amplitude sensitive discriminator comprising a source of alternating current of carrier frequency, a first transformer having its primary of inductance L1 connected to said source with a capacitance C1 across said primary and resonant therewith at the carrier frequency, the secondary of said first transformer being connected to the primary of a second transformer having a secondary of inductance L1 with a centertap, a pair of equal sensing capacitances in series across said centertapped secondary and each having a value of 2C1, means for differentially varying said sensing capacitances in response to the condition being sensed, a signal output transformer having a primary of inductance value L2 equal to tone-quarter of L1 and connected from a point between said series sensing capacitances to said centertap, the primary of said signal output transformer being resonant at the carrier frequency with said two sensing capacitances in parallel with one another, a receiver transformer having its primary connected to the secondary of the signal transformer, the secondary of the receiver transformer having an inductance of Value L2 and being resonant at the carrier frequency with a capacitance of value C2 shunted thereacross and equal to 4C1, first amplifier means connected to the secondary of the receiver transformer and having a first pair of output terminals, a phase and amplitude sensitive discriminator including eight capacitances in series in a loop, said first pair of output terminals being connected to a first pair of Iopposite points in said loop with four of said eight capacitances therebetween on each side said loop, second amplifier means connected to said source and having a second pair of output terminals, said second pair of output terminals being connected to a second pair of opposite points in said loop mid-way between said first pair of opposite points, a rectifier matrix of eight rectification means connected to said loop at four places intermediate said four points, said capacitance loop being resonant with the portion of said first amplifier means between said first pair of output terminals and being resonant with the portion of said second amplifier means between said second pair of output terminals, and an output circuit connected to said rectifier matrix.

2. A measuring system as claimed in claim 1 and wherein the mutual coupling between the primary and secondary of said first transformer equals the mutual coupling between the primary and secondary of said second transformer and wherein the mutual coupling between the primary and secondary of the signal output transformer equals the mutual coupling between the primary and secondary of said receiver transformer.

3. A measurement system including a biresonant capacitance bridge comprising a source of alternating current =of carrier frequency, a first transformer having its primary of inductance L1 connected to said source with a capacitance C1 across said primary and resonant therewith at the carrier frequency, the secondary of said first transformer being connected to the primary of a second transformer having a secondary of inductance L1 with a centertap, a pair of equal sensing capacitances in series across said centertapped secondary'and each having a value lof 2C1, sensing means for differentially varying said sensing capacitances in response to the condition being sensed, a signal output transformer having a primary of inductance value L2 equal to one-quarter of L1 and connected from the centertap to a point between said pair of sensing capacitances, the primary of said signal output transformer being resonant at the carrier frequency with said two sensing capacitances in parallel with each other, a receiver transformer having its primary connected to the secondary of the signal transformer, the secondary of the receiver transformer having an inductance of value L2 and being resonant at the carrier frequency with a capacitance of value C2 shunted thereacross and equal to 4C1, and phase and amplitude sensitive means connected to the secondary of said receiver transformer and to said A.C. source.

4. A measuring system as claimed in claim 3 and wherein the coefficient of coupling between the primary and secondary of each of said four transformers equals the critical coefficient of coupling.

5. A measurement system including a biresonant capacitance bridge and a resonant phase and amplitude sensitive discriminator comprising a source of alternating current of carrier frequency, a first transformer having its primary of inductance L1 connected to said source with a capacitance value C1 across said primary and resonant therewith at the carrier frequency, the secondary of said first transformer being connected to the primary of a second transformer having a secondary of inductance value L1 with a centertap therein, a pair of equal sensing capacitances in series across said centertapped secondary and each having a value of 2C1, sensing means for differentially varying said pair of sensing capacitances in response to the condition being sensed, a signal output transformer having a primary of inductance value L2 equal to one-quarter of L1 and connected from the centertap to a point between said pair lof sensing capacitances, the primary of said signal transformer being resonant at the carrier frequency with said pair of sensing capacitances in parallel, a receiver transformer having its primary connected to the secondary of the signal transformer, the secondary of the receiver transformer having an inductance of value L2 resonant at the carrier frequency with a capacitance of value C2 shunted thereacross and equal to 4C1, first amplifier means connected to the secondary of the receiver transformer and having a first pair of output terminals, a phase and amplitude sensitive discriminator comprising eight capacitances in series in a loop, said pair of output terminals being connected to a first pair of opposite points in said loop with four of said eight capacitances between said first pair of points on opposite sides of the loop, second amplifier means connected to said source and having a second pair of output terminals, said second pair of output terminals being connected to a second pair of opposite points in said loop mid-Way between said first pair of points, and a rectifier matrix connected to said loop at four places intermediate respective adjacent ones of said points, said first and second amplifier means each delivering contraphase voltages at their respective pairs of output terminals and each being resonant with said capacitance loop at the carrier frequency.

6. A measurement system including a biresonant capacitance bridge comprising a source of alternating current of carrier frequency, a first transformer having its primary connected to said source, said primary having a first inductance `with a capacitance across said primary and resonant therewith at the carrier frequency, said first transformer being link coupled to a second transformer having a centertapped secondary of inductance equal to said first inductance and resonant at the carrier frequency with a pair of equal sensing capacitances connected in series across said centertapped'secondary and each having a capacitance value double said first capacitance, sensing CFI means for differentially varying said pair of sensing capacitances as a function of the condition being measured, a signal output transformer having a primary connected from the centertap of said centertapped secondary to a point between said series connected sensing capacitances, said primary of the signal output transformer being resonant with said sensing capacitances in parallel relationship at said carrier frequency, a receiver transformer link coupled to said signal transformer and having a secondary inductance equal to the primary inductance of said signal transformer and resonant at the carrier frequency with a capacitance shunted thereacross, phase and amplitude sensitive discriminator means, and circuit means connecting said discriminator means to the secondary of said receiver transformer and also to said source.

7. A measuring system as claimed in claim 6 and wherein the mutual coupling between the primary and secondary of said first transformer equals the mutual coupling between the primary and secondary of said second transformer and wherein t-he mutual coupling of the primary and secondary of said signal output transformer equals the mutual coupling between the primary and secondary of the receiver transformer.

8. A measurement system including a biresonant capacitance bridge remote from the remainder of the system, said system comprising a source of alternating current of carrier frequency, a first transformer having its primary connected to said source, said primary having a first inductance with a capacitance across said primary and resonant therewith at the carrier frequency, said first transformer being link coupled to a second transformer remote from said first transformer, said second transformer having a centertapped secondary of inductance equal to said first inductance and resonant at the carrier frequency with g a pair of equal sensing capacitances connected in series directly across said centertapped secondary and each having a capacitance value double said first capacitance, sensing means for differentially varying said pair of sensing capacitances as a function of the condition being measured, a signal output transformer having a primary connected from the centertap of said centertapped secondary to a point between said series connected sensing capacitances, said primary of the signal output transformer being resonant with said sensing capacitances in parallel relationship at said carrier frequency, said second transformer, said sensing capacitances and said signal output transformer forming a sensing unit including said biresonant bridge and being remote from the remainder of the system, a receiver transformer remote from said unit and link coupled to said signal transformer and having a secondary inductance equal to the primary inductance of said signal transformer and resonant at the carrier frequency with a capacitance shunted thereacross, phase and amplitude sensitive discriminator means, and circuit means connecting said discriminator means to the secondary of said receiver transformer and also to said source.

1 9. A measuring system including a source of alternating current of carrier frequency, a first transformer having its primary coupled to and energized by said source, the secondary of said first transformer having a centertap, a pair of equal sensing capacitances connected in series across said centertapped secondary, said secondary being resonant at the carrier frequency with said series-connected sensing capacitances, condition sensing means for differentially varying said sensing capacitances in accordance with the condition being sensed for providing a modulated carrier signal, a signal output transformer having a primary of inductance equal to one-quarter of the inductance of the secondaryvof said first transformer and being connected from said centertap to a point between said series connected sensing capacitances, said primary of the signal output transformer being reson-ant at the carrier frequency with said capacitances in parallel with each other, first circuit means connected to the secondary of said signal output transformer and having a pair of output terminals for delivering said modulated carrier signal in contraphase relationship, a phase and amplitude sensitive discriminator :comprising eight capacitances in series in a loop, said pair of output terminals being connected to opposite sides lof said loop, second circuit means coupled to said source and having a second pair of output terminals for delivering the unmodulated carrier signal in contraphase relationship, said second pair of output terminals being connected to opposite sides of said loop midway between said first pair of output terminals, a rectier matrix connected to said loop at four points intermediate said four output terminals, and utilization circuit means connected to said rectifier matrix.

10. A measuring system including a source of alternating current of carrier frequency, a first transformer having its primary coupled to said source and energized thereby, the secondary of said transformer having a centertap, a pair of normally equal sensing capacitances in series directly across said secondary, said secondary being resonant at the carrier frequency with said sensing capacivtances in series, condition sensing means for differentially varying said capacitances in response to the condition being sensed for modulating said carrier signal, an output transformer having a primary of inductance equal to onequarter of the inductance of the secondary of said rst transformer and being connected from the centertap to the mid-point between said pair of sensing capacitances, a phase and amplitude sensitive discriminator, first circuit means connecting said discriminator to the secondary of said signal output transformer for feeding the modulated carrier signal thereto, second circuit means coupling said discriminator to said source for feeding t-he carrier signal thereto, and utilization circuit means connected to said discriminator.

11. A measuring system as claimed in claim and wherein said first and second transformers and said sensing capacitances form a sensing unit remote from the remainder of the system.

12. In a measuring system, a source of alternating current of carrier frequency, a biresonant capacitance condition-sensing bridge comprising a bridge transformer having its primary coupled to and energized by said source, the secondary of said transformer having a centertap forming one of the output terminals of the bridge, a pair of capacitances connected in series across said secondary with a terminal point therebetween forming the other output terminal of the bridge, said pair of capacitances being equal and being resonant in series with the secondary of the bridge transformer at the carrier frequency, vsensing means for varying at least one of said capacitances in response to a condition which is being measured, a signal output transformer having its primary connected to said bridge output terminals and being resonant at the carrier frequency with said pair of capacitances in parallel, phase and amplitude sensitive discriminator means coupled to the secondary of said signal output transformer and to said source, and utilization circuit means connected to said discriminator means.

13. In a measuring system as claimed in claim 12 and having a common return circuit, a biresonant capacitance bridge wherein said terminal point between said pair of capacitances is connected to the common return circuit and said sensing means acts upon the biresonant bridge at said terminal point between the pair of capacitances.

14. In a measuring system as claimed in claim 12, a biresonant capacitance bridge wherein said sensing means differentially varies both of said pair of capacitances in response to the condition being measured.

15. In a measuring system, a source of alternating current of carrier frequency, a biresonant capacitance condition-sensing bridge comprising a bridge transformer having its primary coupled to and energized by said source, the secondary of said transformer having a self inductance L1 and a centertap forming one of the output terminals of the bridge, a pair of capacitances connected in series across said secondary with a terminal point therebetween forming the other output terminal of the bridge, said pair of capacitances being equal and being resonant in series with the secondary of the bridge transformer at the carrier frequency, sensing means for varying at least one of said capacitances in response to a condition which is being measured, a signal output transformer having its primary connected to said bridge output terminals, the primary of said signal output transformer having a self inductance of L2 which is approximately one-quarter of L1, and being resonant at the carrier frequency With said pair of capacitances in parallel, phase and amplitude sensitive discriminator means coupled to the secondary of said signal output transformer and to said source, and utilization circuit means connected to said discriminator means.

16. In a measuring system including a source of alternating current of carrier frequency and a phase and amplitude sensitive discriminator circuit coupled to said source for providing a reference signal to said discriminator circuit, with utilization means connected to said discriminator circuit, a displacement sensing capacitance bridge unit located remotely from said source and discriminator circuit, said unit including a first transformer having a primary connected to said source, a secondary with a centertap, a pair of normally equal capacitors in said unit at least one of which is variable, each capacitor having first and second electrode means, the first electrode means of one capacitor being connected to one side of said secondary, the first electrode means of the other capacitor being connected to the opposite side of said secondary, said second electrode means of both capacitors being conductively connected for connecting said capacitors in series across said secondary, said secondary being resonant with said pair of capacitors in series at said frequency, condition sensing means connected to at least one of said second electrodes for varying the capacitance of at least one capacitor, a second transformer in said unit having a primary with a self inductance one-fourth of the self inductance of said secondary, one side of said latter primary being connected to said centertap and the other side being electrically connected to both of said sec- 0nd electrode means, said latter primary being resonant with said capacitors in parallel at said frequency, the secondary of said second transformer having circuit means connected to said discriminator circuit.

17. In a measuring system including a source of alternating current of carrier frequency and including a phase and amplitude sensitive discriminator circuit coupled to said source for receiving a reference signal for said discriminator circuit, with utilization means connected t0 said discriminator circuit, a displacement sensing capacitance bridge unit located remotely from said source and from said discriminator circuit, said unit including a first transformer having a primary connected to said source, a secondary with a centertap, a pair of normally equal capacitors in said unit, each capacitor having first and second electrode means, the first electrode means of one capacitor being connected to one side of said secondary, the first electrode means of the other capacitor being connected to the opposite side of said secondary, said second electrode means of each capacitor being conductively connected for connecting said capacitors in series across said secondary, said secondary being resonant with said pair of capacitors in series at said frequency, condition sensing means connected to both of said second electrode means for differentially varying the capacitance of both of said capacitors, a second transformer in said unit having a primary with a self inductance approximately one-fourth of the self inductance of said secondary, one side of said latter primary being connected to said centertap and the other side being electrically connected to both of said second electrode means, said latter primary being resonant with said capacitors in parallel at said frequency, the secondary of said second transformer being connected to said discriminator circuit. 

10. A MEASURING SYSTEM INCLUDING A SOURCE OF ALTERNATING CURRENT OF CARRIER FREQUENCY, A FIRST TRANSFORMER HAVING ITS PRIMARY COUPLED TO SAID SOURCE AND ENERGIZED THEREBY, THE SECONDARY OF SAID TRANSFORMER HAVING A CENTERTAP, A PAIR OF NORMALLY EQUAL SENSING CAPACITANCES IN SERIES DIRECTLY ACROSS SAID SECONDARY, SAID SECONDARY BEING RESONANT AT THE CARRIER FREQUENCY WITH SAID SENSING CAPACITANCES IN SERIES, CONDITION SENSING MEANS FOR DIFFERENTIALLY VARYING SAID CAPACITANCES IN RESPONSE TO THE CONDITION BEING SENSED FOR MODULATING SAID CARRIER SIGNAL, AN OUTPUT TRANSFORMER HAVING A PRIMARY OF INDUCTANCE EQUAL TO ONEQUARTER OF THE INDUCTANCE OF THE SECONDARY OF SAID FIRST TRANSFORMER AND BEING CONNECTED FROM THE CENTERTAP TO THE 